Radio apparatus having distortion compensating function

ABSTRACT

In a radio apparatus which compensates for non-linear distortion of a transmission power amplifier, a distributor feeds back the output signal of the transmission power amplifier, an error-signal estimating arithmetic unit digitally estimates an error signal, which is ascribable to non-linear distortion of the power amplifier, using a transmit signal and feedback signal, the transmit signal and the error signal are DA-converted by separate DA converters, and a combiner combines the output signals of the DA converters and inputs the combined signal to the transmission power amplifier to perform distortion compensation.

This application is a continuation division of international applicationnumber PCT 99/04036, filed Jul. 28, 1999.

TECHNICAL FIELD

This invention relates to a radio apparatus having a function whichcompensates for non-linear distortion of a transmission power amplifier.

A power amplifier for amplifying a linear modulated signal used inwireless communications is required to have an amplifier with excellentlinearity in order to suppress deterioration of transmissioncharacteristics caused by spectrum characteristics and signaldistortion. On the other hand, it is required in almost all applicationsthat an amplifier deliver a high power efficiently at all times. Ingeneral, linearity and efficiency of an amplifier are characteristicsthat run counter to each other and a variety of distortion compensationschemes have been proposed in order to reconcile the two.

The field of next-generation mobile telephone systems (IMT-2000, etc.)based upon W-CDMA is one in which the present invention is particularlyuseful. With W-CDMA, code division multiplexing is used indirect-sequence spread-spectrum modulation and multiple-access forsignal modulation. The transmitted signal has a wider band and a higherdynamic range in comparison with the narrow-band modulation and timedivision multiplexing schemes used heretofore in existingsecond-generation mobile telephones (PDC), etc. Accordingly, a poweramplifier used in a W-CDMA apparatus is required to exhibit betterlinearity and higher efficiency than in the past.

BACKGROUND ART

FIG. 29 is a block diagram illustrating an example of a radio apparatusaccording to the prior art. A transmit-signal generator 1 transmits aserial digital data sequence and a serial/parallel (S/P) converter 2divides the digital data sequence alternately one bit at a time toconvert the data to two sequences, namely an in-phase component signal(“I signal”: In-phase component) and a quadrature component signal (“Qsignal”: Quadrature component). A DA converter 3 converts the I and Qsignals to respective ones of analog baseband signals and inputs theseto a quadrature modulator 4. The latter multiplies the input I and Qsignals (the transmit baseband signals) by a reference carrier wave anda signal that has been phase-shifted relative to the reference carrierby 90°, respectively, and sums the results of multiplication to therebyperform quadrature modulation and output the modulated signal. Afrequency converter 5 mixes the quadrature-modulated signal and a localoscillation signal to thereby effect a frequency conversion, and atransmission power amplifier 6 power-amplifies the carrier output fromthe frequency converter 5. The amplified signal is released into theatmosphere from an antenna 7.

In a transmitting apparatus of this kind, the input/outputcharacteristic of the transmission power amplifier developsnon-linearity, as indicated by the dashed line in FIG. 30(a). Owing tothis non-linear characteristic, non-linear distortion occurs and thefrequency spectrum in the vicinity of transmission frequency f₀ developsrising side lobes as indicated by the dashed lines in FIG. 30(b). Thisleads to leakage and interference between neighboring channels. For thisreason, various distortion compensating techniques have been proposed,one of which is a predistorter (an distortion compensating device). Apredistorter adds a characteristic that is the inverse of the distortionof a transmission power amplifier onto an input signal in advance,whereby the transmission power amplifier outputs the desireddistortion-free signal.

FIG. 31 is a block diagram of a radio apparatus having a non-lineardistortion compensating function, which uses a digital Cartesian scheme,as a prior-art example of a predistorter. Digital data sent from thetransmit-signal generator 1 is converted to two signal sequences, namelyan I signal ν_(i) and a Q signal ν_(q), in the S/P converter 2, andthese signals enter a predistorter 8. The predistorter 8 readsdistortion compensation values Δν_(i)(n), Δν_(q)(n), which correspond tothe input baseband signals ν_(i), ν_(q), out of distortion compensationtables 8 a, 8 b, adds these compensation values to the signals ν_(i),ν_(q) and inputs the results to the DA converter 3. The latter convertsthe entered I signal ν_(i) and Q signal ν_(q) to analog baseband signalsand inputs these signals to the quadrature modulator 4. The lattermultiplies the input I and Q signals by a reference carrier wave and asignal that has been phase-shifted relative to the reference carrier by90°, respectively, and sums the results of multiplication to therebyperform quadrature modulation and output the modulated signal. Thefrequency converter 5 mixes the quadrature-modulated signal and a localoscillation signal to thereby effect a frequency conversion, and thetransmission power amplifier 6 power-amplifies the carrier output fromthe frequency converter 5. The amplified signal is released into theatmosphere from the antenna 7. Part of the transmit signal is input to afrequency converter 10 via a directional coupler 9, whereby the signalundergoes a frequency conversion and is input to a quadrature detector11.

The quadrature detector 11 multiplies the input signal by a referencecarrier wave and a signal that has been phase-shifted relative to thereference carrier by 90°, reproduces baseband signals ν′_(i), ν′_(q) onthe transmitting side and applies these signals to an AD converter 12.The latter converts the applied I and Q signals to digital data andinputs the digital data to a distortion compensator 8. At this time aphase shifter 13 applies a phase adjustment in such a manner that thephases of the demodulated baseband signals ν_(i)′, ν_(q)′ will coincidewith the phases of the input signals ν_(i), ν_(q). The AD demodulator 12applies an AD conversion to the demodulated baseband signals ν_(i)′,ν_(q)′ obtained by quadrature detection and inputs the resulting signalsto the predistorter 8. The latter compares the demodulated basebandsignals ν_(i)′, ν_(q)′ and the input baseband signals ν_(i), ν_(q),updates the compensation values in the distortion compensation tables 8a, 8 b based upon errors between the signals and stores updateddistortion compensation values Δν_(i)(n+1), Δν_(q)(n+1) in the memories8 a, 8 b. The operation described above is subsequently repeated.

With the digital Cartesian scheme described above, predistortion iscarried out by obtaining distortion of the transmission power amplifieras an error along each axis of a rectangular coordinate system andadding characteristics that are the inverse of these errors to therespective axial components.

FIG. 32 is a prior-art example of distortion compensation based upon afeed-forward (FF) scheme. With the FF scheme, part of a signal that hasbeen amplified by a main amplifier (transmission power amplifier) 6 isbranched by a directional coupler 9, and an arithmetic unit 15calculates the difference between the branched part of the signal and asignal obtained by subjecting the input signal to a delay adjustment andlevel adjustment. The difference signal is a non-linear distortioncomponent produced by the main amplifier 6. The difference signal isamplified by a linear auxiliary amplifier 16, and a combiner 18combines, 180° out of phase, the output of the auxiliary amplifier and asignal that is result of delaying the main amplifier output by a delayline 17. As a result, distortion compensation is achieved by cancelingout the distortion components.

Problems of the Prior Art

With the conventional predistorter, the signal that has undergonepredistortion is required to have a wide dynamic range in comparisonwith the dynamic range of the original signal in order to compensate foramplitude distortion of the amplifier (the power transmissionamplifier). This means that a higher bit precision is required for theDA converter that subjects the predistortion signal to a DA conversion.In the case of a power amplifier used in W-CDMA in particular, theoriginal signal is a code-multiplexed signal whose amplitude exhibits alarge fluctuation and, moreover, is a wide-band signal owing todirect-sequence spread-spectrum modulation. With the conventionalpredistortion scheme, therefore, the DA converter requires a high bitprecision and, at the same time, a high conversion speed. If suchrequirements are not met, a problem that results is deterioration of thedistortion compensation characteristic.

Further, in predistortion of a power amplifier used in multicarrierW-CDMA, in which multiple carriers undergo common amplification, the DAconverter is required to have even higher speed and higher bit precisioncapabilities. When application to W-CDMA devices currently developed isconsidered, a problem encountered is that the performance of currentlyexisting DA converters cannot satisfy the requirements of high speed andhigh bit precision.

The aforesaid problems arise not only with regard to DA converters buthold true also for AD converters that sample a feedback signal for thepurpose of updating distortion compensation coefficients.

Further, in a radio apparatus having a predistorter that compensates foramplifier distortion as a function of input power, the quadraturemodulator and quadrature demodulator are implemented by analog circuits.A problem encountered is that amplifier-distortion estimation errorgrows owing to imperfections with these quadraturemodulator/demodulators and a satisfactory distortion compensationcharacteristic is not obtained.

With the FF scheme, a problem that arises is that efficiency of theoverall distortion compensating device declines because it is necessaryto use a low-efficiency auxiliary amplifier that requires a high degreeof linearity and because the delay lines and coupler are lossy.

Accordingly, an object of the present invention is to provide a radioapparatus that makes it possible to compensate for distortion of atransmission power amplifier by predistortion even though a DA converterand an AD converter are not required to have high speed and a high bitprecision.

Another object of the present invention is to provide a radio apparatususing a digital quadrature modulator and digital quadrature demodulatorto eliminate the imperfections of analog quadraturemodulator/demodulators, whereby a satisfactory distortion compensationcharacteristic is obtained.

Another object of the present invention is to provide a radio apparatushaving a highly efficient distortion compensation device without use ofan auxiliary amplifier or delay lines.

DISCLOSURE OF THE INVENTION

In a radio apparatus according to the present invention, a transmitsignal (main signal) and a distortion component (error signal) addedonto the main signal are each subjected to a DA conversionindependently, after which the converted signals are combined and inputto a transmission power amplifier. If this arrangement is adopted, theamplitude of the error signal will be small with respect to theamplitude of a predistortion signal obtained by adding a characteristicthat is the inverse of amplitude distortion to the main signal. As aresult, it is possible to lower the bit precision of a DA converter,which outputs only the error signal. Further, a DA converter thatoutputs only the main signal need not have a wide dynamic range and thebit precision of this DA converter an be lowered as well.

Further, in a case where compensation is applied to non-lineardistortion of a transmission power amplifier that amplifies andtransmits a multicarrier signal carrying a plurality of transmitsignals, a signal obtained by DA-converting an error signal is combinedwith a frequency-multiplexed signal obtained by subjecting DA-convertedsignals of respective ones of transmit signals to a frequency-shiftoperation decided by carrier spacing and multiplexing thefrequency-shifted signals, and the combined signal is input to thetransmission power amplifier. If this arrangement is adopted, each ofthe transmit signals and the error signal are DA-converted independentlyand combined. As a result, neither of the DA converters need have a widedynamic range and, hence, bit precision can be suppressed.

In a radio apparatus according to the present invention, predistortionprocessing, in which distortion compensation coefficients are read outof a distortion compensation coefficient table and a transmit signal issubjected to distortion compensation using the distortion compensationcoefficients, and coefficient update processing, in which the distortioncompensation coefficient table is updated using a feedback value(amplifier output signal) that has been sampled by an AD converter, areexecuted separately in terms of time. By thus executing the updating ofdistortion compensation coefficients and predistortion processingseparately in terms of time, a real-time feedback loop is not formed. Asa result, sampled values that are continuous in time are not required asvalues sampled by the AD converter, thus making it possible to mitigatethe requirement that the AD converter have a high speed. Further, sincedistortion compensation is performed by predistortion processing, thereis no need for an auxiliary amplifier and delay lines, which were animpediment to an improvement in efficiency with the FF scheme. Thismakes it possible to raise the overall efficiency the transmission poweramplifier that undergoes distortion compensation.

Further, in a radio apparatus according to the present invention, awide-band sample-and-hold circuit is provided on the input side of an ADconverter that samples a feedback value. If such a wide-bandsample-and-hold circuit is connected to the input of an AD converter anda wide-band signal whose spectrum has spread owing to non-lineardistortion is sampled by this circuit, then it will be possible toperform an AD conversion at a sampling rate lower than the Nyquist rate.In other words, though it is necessary that a sampled band be widenedenough (beyond the Nyquist rate) to enable observation of the distortedsignal, the sampling rate (the number of samples per unit time) can beset independently of the Nyquist rate and the requirement that the ADconverter have a high speed can be mitigated.

In a radio apparatus according to the present invention, a quadraturemodulator and a quadrature demodulator are implemented by digitaloperations. Adopting a digital quadrature modulator/demodulator makes itpossible to reduce the error of the quadrature modulator/demodulator toless than 1 LSB of DA and AD converters. This makes it possible toeliminate deterioration of distortion compensation characteristicscaused by imperfections in a quadrature modulator/demodulator.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a radio apparatus according to a first embodiment of thepresent invention;

FIG. 2 shows an embodiment of an error-signal estimating arithmetic unit(Cartesian scheme);

FIG. 3 shows an embodiment of an error-signal estimating arithmetic unit(polar-loop scheme);

FIG. 4 shows an embodiment of an error-signal estimating arithmetic unit(scheme based upon an LMS algorithm);

FIG. 5 shows an embodiment of an error-signal estimating arithmetic unit(example in which a past transmit signal value is taken into account);

FIG. 6 shows a second embodiment for a case where a transmit signal isprocessed upon being converted to quadrature signals (use of analogquadrature modulator/demodulator);

FIG. 7 shows a third embodiment for a case where a transmit signal isprocessed upon being converted to quadrature signals (use of digitalquadrature modulator/demodulator);

FIG. 8 is a diagram showing the principles of a digital quadraturemodulator;

FIG. 9 is a diagram showing the principles of a digital quadraturedemodulator;

FIG. 10 shows a radio apparatus according to a fourth embodiment of thepresent invention;

FIG. 11 is a diagram showing the structure of an error-signal estimatingarithmetic unit that applies a delay adjustment;

FIG. 12 shows a radio apparatus according to a fifth embodiment of thepresent invention;

FIG. 13 is a diagram showing the structure of a radio apparatusaccording to a sixth embodiment of the present invention, thisembodiment being for a case where multiple transmit signals aretransmitted using a multicarrier signal;

FIG. 14 is a diagram useful in describing a frequency conversion;

FIG. 15 shows a seventh embodiment for a case where a transmit signal isprocessed upon being converted to quadrature signals (use of analogquadrature modulator/demodulator);

FIG. 16 shows an eighth embodiment for a case where a transmit signal isprocessed upon being converted to quadrature signals (use of digitalquadrature modulator/demodulator);

FIG. 17 shows a ninth embodiment for a case where a transmit signal isprocessed upon being converted to quadrature signals (use ofsample-and-hold circuits and analog quadrature modulator/demodulator);

FIG. 18 shows a tenth embodiment for a case where a transmit signal isprocessed upon being converted to quadrature signals (use of asample-and-hold circuit and digital quadrature modulator/demodulator);

FIG. 19 shows an 11^(th) embodiment of a radio apparatus having afunction for correcting amplitude and phase of a reference signal;

FIG. 20 is a diagram showing the structure of a correction circuit;

FIG. 21 is a diagram showing the structure of a correction circuitapplicable to the radio apparatus of the seventh embodiment;

FIG. 22 is a diagram showing the structure of a correction circuitapplicable to the radio apparatus of the eighth embodiment;

FIG. 23 shows a radio apparatus according to a 12th embodiment of thepresent invention;

FIG. 24 shows a radio apparatus according to a 13th embodiment of thepresent invention;

FIG. 25 shows a radio apparatus according to a 14th embodiment of thepresent invention;

FIG. 26 shows a radio apparatus according to a 15th embodiment of thepresent invention;

FIG. 27 is a diagram showing the principles of a sampling-rateconverter;

FIG. 28 is a diagram showing the principles of a sampling-rateconverter;

FIG. 29 is a diagram showing the structure of a transmitting apparatusaccording to the prior art;

FIG. 30 is a diagram useful in describing problems ascribable tonon-linearity of a transmission power amplifier;

FIG. 31 shows the structure (Cartesian scheme) of a transmittingapparatus having a digital non-linear distortion compensating functionaccording to the prior art; and

FIG. 32 is a diagram showing the structure of a prior-art example of afeed-forward system.

BEST MODE FOR CARRYING OUT THE INVENTION (A) First Embodiment

FIG. 1 is a diagram showing the structure of a first embodiment of thepresent invention. A digital transmit signal x(t) output from atransmit-signal generator (not shown) is converted to an analog signal(main signal S_(M)) by a DA converter 31, and a combiner 32 combinesthis signal with an error signal S_(E), which is output from anerror-signal estimating arithmetic unit 33, and inputs the combinedsignal to a power amplifier (amp) 34. Part of the amplifier output isbranched by a distributor 35 such as a directional coupler and is thensampled by an AD converter 36, thus becoming a feedback signal S_(F).The error-signal estimating arithmetic unit 33 uses the transmit signalx(t) and the feedback signal S_(F) to calculate an error ascribable tonon-linear distortion of the amplifier. A DA converter 37 subjects theobtained error signal to a DA conversion, an attenuator 38 adjusts thelevel of the error signal and then the combiner 32 combines the mainsignal S_(M) and the error signal S_(E) and inputs the combined signalto the amplifier. It should be noted that it is also possible to adoptan arrangement in which the combined signal output from the combiner 32is input to the amplifier 34 after bring frequency-converted to an RFsignal (IF→RF) and the amplifier output signal from the distributor 35is input to the AD converter 36 after being frequency-converted to an IFsignal (RF→IF).

Thus, the amplifier input signal is a signal obtained by adding on acharacteristic that is the inverse of the non-linear distortion of theamplifier (a signal that has been subjected to predistortion), andtherefore a linear amplified output signal is obtained at the amplifieroutput.

FIG. 2 shows a first embodiment of the error-signal estimatingarithmetic unit. This is an example using a predistorter based upon adigital Cartesian scheme. The error-signal estimating arithmetic unit 33calculates error signals (ν_(err,i), ν_(err,q)) as equivalent basebandsignals (complex numbers) using a rectangular coordinate system. Errorsignals conforming to the respective components of a transmit basebandsignal (x_(i),x_(q)) have been stored in memory tables 33 a, 33 b,respectively.

When the transmit signal x(t) is generated, the error signals(ν_(err,i), ν_(err,q)) conforming to the transmit baseband signals(x_(i),x_(q)) are read out of the memory tables 33 a, 33 b and undergovector addition with the transmit signals (x_(i),x_(q)) in the combiner32 (FIG. 1), whereby predistortion is performed. Arithmetic units 33 c,33 d calculate difference vectors (x_(i)′-x_(i),x_(q)′-x_(q)) betweenthe transmit baseband signals (x_(i),x_(q)) and feedback signals(x_(i)′,x_(q)′), multipliers 33 e, 33 f multiply the difference vectorsby a step-size parameter μ (0<μ<1), and adders 33 g, 33 h add themultiplier outputs to the error signals (ν_(err,i), ν_(err,q)) and storethe results of addition at memory addresses corresponding to the axialcomponents of the transmit baseband signals (x_(i),x_(q)), therebyachieving updating.

FIG. 3 is a second embodiment of the error-signal estimating arithmeticunit. This is an example using a predistorter based upon a polar-loopscheme. The error-signal estimating arithmetic unit 33 calculates theerror signals (ν_(err,i), ν_(err,q)) in an equivalent baseband systemusing a polar coordinate system. Error signals (r_(err),φ_(err))conforming to (r,φ), which are the result of subjecting the transmitbaseband signals (x_(i),x_(q)) to a polar transformation, have beenstored in memory tables 33 a, 33 b, respectively.

If the transmit baseband signals (x_(i),x_(q)) are input, arectangular-to-polar coordinate transformation circuit 33 i transformsthe transmit baseband signals to polar coordinates (γ,φ) and reads errorsignals (r_(err),φ_(err)), which conform to the transmit signals (r,φ),out of the memory tables 33 a, 33 b. A polar-to-rectangular coordinatetransformation circuit 33 j transforms the error signals(r_(err),φ_(err)) to the error signals (ν_(err,i), ν_(err,q)) in therectangular coordinate system, and the combiner 32 (FIG. 1) performspredistortion by performing vector addition between the error signals(ν_(err,i), ν_(err,q)) and the transmit baseband signals (x_(i),x_(q)).

A rectangular-to-polar coordinate transformation circuit 33 k transformsthe feedback signals (x_(i)′,x_(q)′) to polar coordinates (γ′,φ′). Thearithmetic units 33 c, 33 d calculate difference vectors (r′-r,φ′-φ)between the transmit baseband signals (r,φ) and feedback signals(r′,φ′), the multipliers 33 e, 33 f multiply the difference vectors by astep-size parameter μ (0<μ<1), and the adders 33 g, 33 h add themultiplier outputs to the error signals (r_(err),φ_(err)) and store theresults of addition at memory addresses corresponding to the componentsof the transmit signals (r,φ), thereby achieving updating.

FIG. 4 is a third embodiment of the error-signal estimating arithmeticunit. This is an example using an adaptive predistorter. An adaptivepredistorter is a scheme for performing predistortion by estimatingamplifier distortion using an LMS adaptive algorithm and adding acharacteristic that is the inverse of this distortion to a transmitsignal by complex multiplication.

The error-signal estimating arithmetic unit 33 has a multiplier 33 m formultiplying the transmit signal x(t) by a distortion compensationcoefficient h_(n−1)(p), and an arithmetic unit 33n for subtractingh_(n−1)*x(t) from the transmit signal x(t) and outputting an errorsignal ν_(err) [=h_(n)(p)x(t)-x(t)]. An arithmetic unit 33 p calculatespower p [=x(t)²] of the transmit signal x(t), and a distortioncompensation coefficient storage unit 33 q stores a distortioncompensation coefficient h(p) that conforms to each power of thetransmit signal x(t) and outputs a distortion compensation coefficienth_(n−1)(p) conforming to the power p of the transmit signal x(t).Further, the distortion compensation coefficient storage unit 33 qupdates the distortion compensation coefficient h_(n−1)(p) by adistortion compensation coefficient h_(n)(p) decided by the LMSalgorithm.

Reference characters 33 r represent a complex-conjugate signal outputunit, 33 s a subtractor for outputting a difference e(t) between thetransmit signal x(t) and a feedback demodulation signal y(t), 33 t amultiplier for multiplying e(t) and u*(t), 33 u a multiplier formultiplying h_((n−1))(p) and y*(t) and outputting u*(t), 33 v amultiplier for multiplying by a step-size parameter, and 33 w an adderfor adding h_((n−1))(p) and μe(t)u*(t). An operation in accordance withthe following LMS algorithm is performed by the arrangement set forthabove: The arithmetic operations performed by the arrangement set forthabove are as follows:

h _(n)(p)=h _(n−1)(p)+μe(t)u*(t)  (1)

e(t)=x(t)−y(t)

y(t)=h_(n−1)(p)x(t)f(p)

u(t)=x(t)f(p)≈h* _(n−1)(p)y(t)h _(n−1)(p)h* _(n−1)(p)≈1

p=|x(t)|²

where

x(t): input baseband signal (transmit signal)

f(p): amplifier distortion function

h(p): estimated distortion compensation coefficient

μ: step-size parameter

y(t): feedback signal

u(t): distorted signal

Here x, y, f, h, u, e represent complex numbers and signifies a complexconjugate. The signal u(t) approximates [(h_(n−1)(p)·h*_(n−1)(p)≈1] ifit is assumed that amplitude distortion of the amplifier is not verylarge.

By executing the processing set forth above, the distortion compensationcoefficient h(p) is updated in accordance with Equation (1) so as tominimize the difference e(t) between the transmit signal x(t) and thefeedback signal y(t), and the coefficient eventually converges to theoptimum distortion compensation coefficient so that compensation is madefor the distortion in the transmission power amplifier.

FIG. 5 is a fourth embodiment of the error-signal estimating arithmeticunit. This is an example using an adaptive predistorter, in whichcomponents identical with those of the embodiment of FIG. 4 aredesignated by like reference characters. This embodiment differs in that(1) distortion compensation coefficients are stored and updated in thedistortion compensation coefficient storage unit 33 q in correspondencewith combinations of present momentary power P=|x(t)|² and a functiong(x) of present and past inputs, and (2) an arithmetic unit 33 x forcalculating the function g(x) is provided. FIG. 5 illustrates an examplein which a difference Δp between present power and previous power istaken as the function g(x) [Δp=|x(t)|²−|x(t−1)|²].

The predistorter of FIG. 5 treats the amplifier as a distortiontransmission path having a memory in order to compensate for frequencyasymmetric distortion of the amplifier and performs predistortion byestimating this distortion using the LMS adaptive algorithm and adding acharacteristic that is the inverse of this distortion to a transmitsignal by complex multiplication. In order to compensate for distortionthat has been influenced by past input amplitude, distortioncompensation coefficients are stored and updated as a table regardingthe two dimensions of present momentary power P=|x(t)|² and the functiong(x) (=Δp) of present and past inputs.

The error signal ν_(err)(t)=h*x(t)−x(t) is obtained by subtracting thetransmit signal x(t) from the transmit signal h_(n−1)·x(t), which hasundergone distortion compensation, using the adapting predistorter ofthis embodiment.

(B) Second Embodiment

FIG. 6 shows a second embodiment for a case where the transmit signalx(t) is processed upon being converted to quadrature signals. Componentsidentical with those of the first embodiment of FIG. 1 are designated bylike reference characters. The second embodiment differs from the firstembodiment in that the transmit signal, error signal and feedback signalare calculated in an equivalent baseband system (complex coordinatesystem). To achieve this, an S/P converter 41 for dividing the transmitsignal x(t) alternately one bit at a time to convert the signal toquadrature signals (I and Q signals), quadrature modulators 42, 43 and aquadrature demodulator 44 are provided. Further, DA converters 31 a, 31b are provided as the DA converter 31 for DA-converting the componentsof the quadrature signals, DA converters 37 a, 37 b are provided as theDA converter 37 for DA-converting the components of the error signal,and AD converters 36 a, 36 b are provided as the AD converter 36 forAD-converting the in-phase and quadrature components of the feedbacksignal.

The S/P converter 41 converts the transmit signal x(t) to quadraturesignals and the DA converters 31 a, 31 b convert the quadrature-signalcomponents to analog signals and input the analog signals to thequadrature modulator 42. The latter generates the main signal S_(M) byapplying quadrature modulation to the in-phase component and quadraturecomponent (I and Q signals) input from the DA converters 31 a, 31 b. Thecombiner 32 combines the main signal S_(M) with the error signal S_(E),which is output from the error-signal estimating arithmetic unit 33, andinputs the combined signal to the power amplifier (amp) 34. Part of theamplifier output is branched by the distributor 35, such as thedirectional coupler, and enters the quadrature demodulator 44. Thelatter reconstructs the baseband signals on the transmit side byapplying quadrature demodulation to the input signal and inputs the I, Qsignals to the AD converters 36 a, 36 b. These AD converters subject theI, Q signals to an AD conversion and input the results to theerror-signal estimating arithmetic unit 33 as the feedback signal y(t).

The error-signal estimating arithmetic unit 33 uses the transmit signalx(t) and the feedback signal y(t) to calculate an error signalascribable to non-linear distortion of the amplifier. The DA converters37 a, 37 b subject the in-phase component and quadrature component,respectively, of the obtained error signal to a DA conversion and inputthe results to the quadrature modulator 43. The latter generates theerror signal S_(E) by applying quadrature modulation to the in-phasecomponent and quadrature component (I and Q signals) that enter from theDA converters 37 a, 27 b. The attenuator 38 adjusts the level of theerror signal and then the combiner 32 combines the main signal S_(M) andthe error signal S_(E) and inputs the combined signal to the amplifier.

It should be noted that it is also possible to adopt an arrangement inwhich the combined signal output from the combiner 32 is input to theamplifier 34 after bring frequency-converted to an RF signal (IF→RF) andthe amplifier output signal from the distributor 35 is input to the ADconverter 36 after being frequency-converted to an IF signal (RF→IF).

Thus, the amplifier input signal is a signal obtained by adding on acharacteristic that is the inverse of the non-linear distortion of theamplifier (a signal that has been subjected to predistortion), andtherefore a linear amplified output signal is obtained at the amplifieroutput.

(C) Third Embodiment

FIG. 7 shows a third embodiment for a case where the transmit signal isprocessed upon being converted to quadrature signals. Componentsidentical with those of the first embodiment of FIG. 1 are designated bylike reference characters. The third embodiment differs from the firstembodiment in that the transmit signal, error signal and feedback signalare calculated in an equivalent baseband system (complex coordinatesystem), and in that there are provided the S/P converter 41 fordividing the transmit signal x(t) alternately one bit at a time toconvert the signal to quadrature signals (I and Q signals), digitalquadrature modulators 45, 46 and a digital quadrature demodulator 47.

The S/P converter 41 converts the transmit signal x(t) to quadraturesignals, the digital quadrature modulator 45 applies quadraturemodulation to the in-phase component and quadrature component (I and Qsignals) input from the S/P converter, and the DA converter 31 generatesthe main signal S_(M) by converting the digital quadrature-modulatedsignal to an analog signal. The combiner 32 combines the main signalS_(M) with the error signal S_(E) output from the error-signalestimating arithmetic unit 33 and inputs the combined signal to thepower amplifier (amp) 34. Part of the amplifier output is branched bythe distributor 35, such as the directional coupler, and enters thedigital quadrature demodulator 47. The latter reconstructs the basebandsignals on the transmit side by applying quadrature demodulation to theinput signal and inputs the in-phase component and quadrature componentto the error-signal estimating arithmetic unit 33 as the feedback signaly(t).

The error-signal estimating arithmetic unit 33 uses the transmit signalx(t) and the feedback signal y(t) to calculate an error signalascribable to non-linear distortion of the amplifier. The digitalquadrature modulator 46 applies quadrature modulation to the inputin-phase component and quadrature component (I and Q signals), and theDA converter 37 generates the error signal S_(E) by converting thedigital quadrature-modulated signal (error signal) to an analog signal.The attenuator 38 adjusts the level of the error signal and then thecombiner 32 combines the main signal S_(M) and the error signal S_(E)and inputs the combined signal to the amplifier.

It should be noted that it is also possible to adopt an arrangement inwhich the combined signal output from the combiner 32 is input to theamplifier 34 after bring frequency-converted to an RF signal (IF→RF) andthe amplifier output signal from the distributor 35 is input to the ADconverter 36 after being frequency-converted to an IF signal (RF→IF).

Thus, the amplifier input signal is a signal obtained by adding on acharacteristic that is the inverse of the non-linear distortion of theamplifier (a signal that has been subjected to predistortion), andtherefore a linear amplified output signal is obtained at the amplifieroutput.

FIG. 8 shows an embodiment of a digital quadrature modulator. A signadd-on unit 45 a adds a sign onto an in-phase component ν_(i), and asign add-on unit 45 b adds a sign onto a quadrature component ν_(q). Aselector 45 c selects baseband signals in accordance with the sequenceν_(i)(t), −ν_(q)(t+Ts), −ν_(i)(t+2Ts), ν_(q)(t+3Ts) (where Ts representsthe sampling period), etc., and inputs the baseband signal to the DAconverter 31. The latter converts the input signal to an analog signaland outputs the analog signal as the main signal S_(M). Frequency f_(IF)of the output modulated signal thus obtained and sampling frequency (thechangeover frequency of the selector) f_(sample) of the DA converter 31are related as indicated by the following equation:

f _(sample)=4×f _(IF)  (2)

Digital quadrature modulation is thus carried out to generate themodulated signal.

FIG. 9 shows an embodiment of a digital quadrature demodulator. Areceive IF signal (or RF signal) is sampled by the AD converter 36.Quadrature demodulation is performed by selecting the intermediatefrequency f_(IF) and f_(sample) so as to satisfy the requirementindicated by the following equation:

f _(sample)=4×f _(IF)/(4m+1)  (3)

where m represents a natural number. If the in-phase component andquadrature component of a sampled value obtained in accordance with theabove requirement are v_(i) and ν_(q), respectively, then a selector 47a outputs the input signal in accordance with the sequence ν_(i)(t),−ν_(q)(t+Ts), −ν_(i)(t+2Ts), ν_(q)(t+3Ts), etc. That is, the selector 47a divides the output sample sequence into even-numbered and odd-numberedsamples, inputs the even-numbered samples to a sign reversing unit 47 band inputs the odd-numbered samples to a sign reversing unit 47 c, andthe sign reversing units 47 b, 47 c alternately reverse the signs oftheir input signals, thereby generating quadrature demodulated resultsν_(i), ν_(q). Here ν_(i), ν_(q) are the in-phase and quadraturecomponents, respectively, and are obtained as sample sequences offsetfrom each other in terms of sampling time. Accordingly, sample values ofthe desired timing are found by interpolation and filtering processing.Digital quadrature detection is thus carried out to obtain theequivalent baseband signal (complex signal).

(D) Fourth Embodiment

FIG. 10 is a diagram showing the structure of a fourth embodiment of thepresent invention, in which components identical with those of thesecond embodiment of FIG. 6 are designated by like reference characters.The fourth embodiment differs from the second embodiment in thatsample-and-hold circuits 48 a, 48 b for sampling and holding thein-phase component and quadrature component output from the quadraturedemodulator 44 are provided in front of the AD converters 36 a, 36 b.

The sampling clock f_(sample) of the AD converters 36 a, 36 b and acontrol signal f_(SH) of the sample-and-hold circuits 48 a, 48 b aresupplied independently. The sampling band is decided by the pulse widthof the control signal f_(SH), which controls the sampling rate of thesample-and-hold circuits 48 a, 48 b, and the sampling rate (number ofsamples per unit time) is decided by the sampling clock f_(sample) ofthe AD converters.

The sample sequence obtained is input to the error-signal estimatingarithmetic unit 33 as feedback values and is used to update thedistortion compensation coefficients. As shown in FIG. 11, theerror-signal estimating arithmetic unit 33 can have a structure thatdoes not require the feedback values to be continuous in time. Thismakes it possible to perform down sampling in which the sampling rate isset to be lower than the Nyquist rate, which is decided by the samplingband. It should be noted that the error-signal estimating arithmeticunit 33 of FIG. 11 is obtained by adding a delay circuit DLY to thearrangement of FIG. 4.

The sample-and-hold circuits 48 a, 48 b sample and hold the in-phasecomponent and quadrature component, which is output from the quadraturedemodulator 44, in accordance with the control signal f_(SH), and the ADconverters 36 a, 36 b convert the held signals to digital signals inaccordance with the sampling clock f_(sample), thereby obtaining thefeedback values. The sampling timing of the sample-and-hold circuits 48a, 48 b is decided by the band characteristic of the entered distortedsignal sampled, and the hold timing is decided by the conversion timenecessary to subject the held signal to the AD conversion. Further, therate of the digital signal fed back to the error-signal estimatingarithmetic unit 33 is a rate made to conform to the AD conversion rate.

The error-signal estimating arithmetic unit 33 delays the referencesignal (transmit signal) by the delay line DLY, thereby obtaining areference signal whose time corresponds to that of the feedback signal,calculates a distortion compensation coefficient in accordance with theLMS adaptive algorithm in such a manner that the difference between thedelayed reference signal and feedback signal will become zero, andstores the above-mentioned distortion compensation coefficient at anaddress of the distortion compensation coefficient storage unit 33 qindicated by the delayed power value, thereby achieving updating. Theupdating of the distortion compensation coefficient at this time iscarried out at a rate that conforms to the AD conversion rate.Meanwhile, multiplication (predistortion) of the transmit signal by thedistortion compensation coefficient is performed at the sampling rate ofthe transmit signal using a distortion compensation coefficient that hasbeen stored at an address of the distortion compensation coefficientstorage unit 33 q that corresponds to the undelayed transmit signal.More specifically, predistortion and updating of the distortioncompensation coefficient are executed independently using a dual-portRAM or the like.

In accordance with the fourth embodiment, effective feedback values canbe sampled under the constraint of device performance in distortioncompensation of W-CDMA, which requires a wide band and large dynamicrange, by combining, e.g., wide-band sample-and-hold circuits andhigh-precision AD converters.

(E) Fifth Embodiment

FIG. 12 is a diagram showing the structure of a fifth embodiment, inwhich components identical with those of the third embodiment of FIG. 7are designated by like reference characters. This embodiment differsfrom the third embodiment in that a sample-and-hold circuit 49 forsampling and holding the output signal of the amplifier 34 is providedin front of the AD converter 36.

The sample-and-hold circuit 49 samples and holds the IF signal (or RFsignal) in accordance with the control signal f_(SH), the AD converter36 converts the held signal to a digital signal in accordance with thesampling clock f_(sample), and the digital quadrature demodulator 47applies quadrature demodulation processing to this digital signaldigitally to generate the in-phase component and quadrature component ofthe feedback signal, and inputs these signals to the error-signalestimating arithmetic unit 33. The relationship between the samplingband and the sampling rate and the operation of the error-signalestimating arithmetic unit are similar to those of the fourthembodiment.

In accordance with the fifth embodiment, effective feedback values canbe sampled under the constraint of device performance in distortioncompensation of W-CDMA, which requires a wide band and large dynamicrange, by combining, a wide-band sample-and-hold circuit and ahigh-precision AD converters.

(E) Sixth Embodiment

FIG. 13 is a diagram showing the structure of a radio apparatusaccording to a sixth embodiment of the present invention, thisembodiment being for a case where multiple transmit signals aretransmitted using a multicarrier signal. This illustrates an example fora case where four frequencies are multiplexed and transmitted.Components identical with those of the first embodiment are designatedby like reference characters.

Transmit signals x₁(t), x₂(t), x₃(t) and x₄(t) of respective carriersignals are converted to analog signals by independent DA converters 51₁˜51 ₄, respectively, the analog signals are frequency-converted todesired carrier frequencies f₁, f₂, f₃, f₄ by frequency converters 53₁˜53 ₄ after passage through filters 52 ₁˜52 ₄ [see (a) of FIG. 14], andthe signals are frequency-multiplexed by a combiner 54.

The frequency-multiplexed (main signal) S_(M) obtained is combined inthe combiner 32 with the error signal S_(E) output from the error-signalestimating arithmetic unit 33, and the combined signal is input to thepower amplifier (amp) 34. Part of the amplifier output is branched bythe distributor 35, such as the directional coupler, and isfrequency-converted to a frequency-multiplexed signal of frequenciesf₁-f₀, f₂-f₀, f₃-f₀, f₄-f₀ by a frequency converter 55. This signal isAD-converted by the AD converter 36 after passage through a filter 56and becomes the feedback signal S_(F).

Meanwhile, the digital values of the transmit signals x₁(t), x₂(t),x₃(t), x₄(t) are multiplied by exp(jω₁t), exp(jω₂t), exp(jω₃t),exp(jω₄t) (ω_(n)=2πf_(n)), respectively, by frequency shifters 57 ₁˜57₄, respectively, to effect a frequency shift to frequencies f₁, f₂, f₃,f₄, after which these frequencies are frequency-multiplexed by acombiner 58. This digital frequency-multiplexed signal is a signalequivalent to the main signal S_(M) obtained by combining theabove-mentioned individual carrier signals in analog fashion. Afrequency shifter 59 subsequently multiplies this digitalfrequency-multiplexed signal by exp(−jω₀t) to effect a conversion to thefrequency multiplexed signal of frequencies f₁-f₀, f₂-f₀, f₃-f₀, f₄-f₀[see (b) of FIG. 14] and inputs this signal to the error-signalestimating arithmetic unit 33 as a reference signal S_(R).

An arrangement can also be adopted in which the frequency shifter 59 isdeleted and multiplication by exp[jω₁-ω₀)t], exp[jω₂-ω₀)t],exp[j(ω₃-ω₀)t], exp[j(ω₄-ω₀)t] is performed by the frequency shifters 57₁˜57 ₄. This holds true also for embodiments below. Further, in a casewhere a frequency shift of f₀ is not performed by the frequencyconverter 55, the frequency shifter 59 and a frequency converter 61,described later, will be unnecessary.

The error-signal estimating arithmetic unit 33 uses the reference signalS_(R) and the feedback signal S_(F) to calculate an error signalascribable to non-linear distortion of the amplifier. The DA converter37 DA-converts the obtained error signal and inputs the analog signal tothe frequency converter 61 via a filter 60. The frequency converter 61multiplies the error signal by a signal of frequency f₀ to therebyup-convert the error signal frequency. After the signal has its leveladjusted by the attenuator, it is combined with the main signal. Theattenuator 38 adjusts the level of the error signal and then thecombiner 32 combines the main signal S_(M) and the error signal S_(E)and inputs the combined signal to the amplifier. Thus there is obtaineda signal that is the result of adding a characteristic that is theinverse of the non-linear distortion of the amplifier to thefrequency-multiplexed signal (main signal).

(G) Seventh Embodiment

FIG. 15 shows a seventh embodiment for a case where the transmit signalsare processed upon being converted to quadrature signals. Componentsidentical with those of the sixth embodiment of FIG. 13 are designatedby like reference characters. The seventh embodiment differs from thesixth embodiment in that the transmit signals, error signal and feedbacksignal are calculated in an equivalent baseband system (complexcoordinate system). To accomplish this, there are provided S/Pconverters 71 ₁˜71 ₄ for converting the transmit signals x₁(t)˜x₄(t) toquadrature signals (I and Q signals), quadrature modulators 72 ₁˜72 ₄,73 and a quadrature demodulator 74. Further, DA converters 51 ₁₁, 51₁₂˜51 ₄₁, 51 ₄₂ for DA-converting each of the in-phase and quadraturecomponents of the transmit signals are provided as the DA converters 51₁˜51 ₄, DA converters 37 a, 37 b for DA-converting the in-phase andquadrature components of the error signal are provided as the DAconverter 37, and the AD converters 36 a, 36 b for AD-converting thein-phase and quadrature components of the feedback signal are providedas the AD converter 36.

The S/P converters 71 ₁˜71 ₄ convert the transmit signals x₁(t)˜x₄(t) toquadrature signals, and the DA converters 51 ₁₁, 51 ₁₂˜51 ₄₁, 51 ₄₂convert the in-phase and quadrature components of each of the quadraturesignals to analog signals and input the analog signals to the quadraturemodulators 72 ₁˜72 ₄. The latter apply quadrature modulation to thein-phase and quadrature components (I and Q signals) that enter from thecorresponding DA converters 51 ₁₁, 51 ₁₂˜51 ₄₁, 51 ₄₂, the frequencyconverters 53 ₁˜53 ₄ up-convert the frequencies of thequadrature-modulated signals to the desired carrier frequencies f₁, f₂,f₃, f₄, and the combiner 54 performs frequency multiplexing to generatethe main signal S_(M).

The combiner 32 combines the main signal S_(M) with the error signalS_(E) output from the error-signal estimating arithmetic unit 33 andinputs the combined signal to the power amplifier (amp) 34. Thedistributor 35 branches part of the amplifier output and the frequencyconverter 55 performs a down-conversion to a frequency-multiplexedsignal of frequencies f₁-f₀, f₂-f₀, f₃-f₀, f₄-f₀ and inputs this signalto the quadrature demodulator 74 via the filter 56. The quadraturedemodulator 74 subjects the input signal to quadrature demodulationprocessing to reconstruct the baseband signals on the transmit side, theI, Q signals are input to the AD converters 36 a, 36 b and these ADconverters subject the I, Q signals to an AD conversion and input theresults to the error-signal estimating arithmetic unit 33 as thefeedback signal.

Meanwhile, quadrature signals obtained by subjecting the transmitsignals x₁(t), x₂(t), x₃(t), x₄(t) to a quadrature conversion aremultiplied by exp(jω₁t), exp(jω₂t), exp(jω₃t), exp(jω₄t)(ω_(n)=2πf_(n)), respectively, by the frequency shifters 57 ₁˜57 ₄,respectively, to effect a frequency shift to frequencies f₁, f₂, f₃, f₄,after which these frequencies are frequency-multiplexed by the combiner58. The frequency shifter 59 subsequently multiplies this digitalfrequency-multiplexed signal by exp(−jω₀t) to effect a conversion to thefrequency multiplexed signal of frequencies f₁-f₀, f₂-f₀, f₃-f₀, f₄-f₀and inputs this signal to the error-signal estimating arithmetic unit 33as the reference signal S_(R).

The error-signal estimating arithmetic unit 33 uses the reference signaland the feedback signal to calculate an error signal ascribable tonon-linear distortion of the amplifier. The DA converters 37 a, 37 _(b)DA-convert the in-phase and quadrature components of the obtained errorsignal and input the analog signals to the quadrature modulator 73 viafilters 60 a, 60 b. The quadrature modulator 73 subjects the inputsignals to quadrature modulation and inputs the modulated signal to thefrequency converter 61. The latter multiplies the error signal by thesignal of frequency f₀ to up-convert the error signal frequency. Theattenuator 38 adjusts the level of the error signal and then thecombiner 32 combines the level-adjusted error signal with the mainsignal and inputs the combined signal to the amplifier. Thus there isobtained a frequency-multiplexed signal that is the result of adding ona characteristic that is the inverse of the non-linear distortion of theamplifier.

(H) Eighth Embodiment

FIG. 16 shows an eighth embodiment for a case where the transmit signalsare processed upon being converted to quadrature signals. Componentsidentical with those of the sixth embodiment of FIG. 13 are designatedby like reference characters. The eighth embodiment differs from thesixth embodiment in that the transmit signals, error signal and feedbacksignal are calculated in an equivalent baseband system (complexcoordinate system). The S/P converters 71 ₁˜71 ₄ are provided and so aredigital quadrature modulators 75 ₁˜75 ₄, 76 and a digital quadraturedemodulator 77.

The S/P converters 71 ₁˜71 ₄ convert the transmit signals x₁(t), x₂(t),x₃(t), x₄(t) to quadrature signals (I and Q signals), the digitalquadrature modulators 75 ₁˜75 ₄ apply digital quadrature modulation tothe in-phase and quadrature components (I and Q signals) of each of thequadrature signals, and the DA converters 51 ₁˜51 ₄ convert the digitalquadrature-modulated signals to analog signals and input the analogsignals to the frequency converters 53 ₁˜53 ₄ via the filters. Thefrequency converters 53 ₁˜53 ₄ up-convert the frequencies of thequadrature-modulated signals to the desired carrier frequencies f₁, f₂,f₃, f₄ and the combiner 54 performs frequency multiplexing to generatethe main signal S_(M).

The combiner 32 combines the main signal S_(M) with the error signalS_(E) output from the error-signal estimating arithmetic unit 33 andinputs the combined signal to the power amplifier (amp) 34. Thedistributor 35 branches part of the amplifier output and the frequencyconverter 55 performs a down-conversion to a frequency-multiplexedsignal of frequencies f₁-f₀, f₂-f₀, f₃-f₀, f₄-f₀ and inputs this signalto the AD converter 36 the filter 56. The AD converter 36 converts theinput signal to a digital signal and inputs the digital signal to thedigital quadrature demodulator 47. The latter subjects the input signalto quadrature demodulation processing to reconstruct the basebandsignals on the transmit side, and the in-phase and quadrature componentsare input to the error-signal estimating arithmetic unit 33 as thefeedback signal.

Meanwhile, quadrature signals obtained by subjecting the transmitsignals x₁(t), x₂(t), x₃(t), x₄(t) to a quadrature conversion aremultiplied by exp(jω₁t), exp(jω₂t), exp(jω₃t), exp(jω₄t)(ω_(n)=2πf_(n)), respectively, by the frequency shifters 57 ₁˜57 ₄,respectively, to effect a frequency shift to frequencies f₁, f₂, f₃, f₄,after which these frequencies are frequency-multiplexed by the combiner58. The frequency shifter 59 subsequently multiplies this digitalfrequency-multiplexed signal by exp(−jω₀t) to effect a conversion to thefrequency multiplexed signal of frequencies f₁-f₀, f₂-f₀, f₃-f₀, f₄-f₀and inputs this signal to the error-signal estimating arithmetic unit 33as the reference signal S_(R).

The error-signal estimating arithmetic unit 33 uses the reference signaland the feedback signal to calculate an error signal ascribable tonon-linear distortion of the amplifier and inputs the in-phase componentand quadrature component of this signal to the transmit-data processingunit 76. The latter applies quadrature modulation to the input in-phaseand quadrature components (I and Q signals), and the DA converter 37converts the digital quadrature-modulated signal (error signal) to ananalog signal and inputs the modulated signal to the frequency converter61 via the filter 60. The frequency converter 61 multiplies the errorsignal by the signal of frequency f₀ to up-convert the frequency. Theattenuator 38 adjusts the level of the error signal and the combiner 32combines the level-adjusted error signal with the main signal and inputsthe combined signal to the amplifier. Thus there is obtained afrequency-multiplexed signal that is the result of adding on acharacteristic that is the inverse of the non-linear distortion of theamplifier.

(I) Ninth Embodiment

FIG. 17 is a diagram showing the structure of a ninth embodiment of thepresent invention, in which components identical with those of theseventh embodiment of FIG. 15 are designated by like referencecharacters. This embodiment differs from the seventh embodiment in thatsample-and-hold circuits 78 a, 78 b for sampling and holding thein-phase component and quadrature component output from the quadraturedemodulator 74 are provided in front of the AD converters 36 a, 36 b.The operation of the sample-and-hold circuits 78 a, 78 b, AD converters36 a, 36 b and error-signal estimating arithmetic unit 33 is identicalwith that of the fourth embodiment of FIG. 10 and the same effects areobtained.

(J) Tenth Embodiment

FIG. 18 is a diagram showing the structure of a tenth embodiment of thepresent invention, in which components identical with those of theeighth embodiment of FIG. 16 are designated by like referencecharacters. This embodiment differs from the eighth embodiment in that asample-and-hold circuit 79 for sampling and holding the output signal ofthe amplifier 34 is provided in front of the AD converter 36. Theoperation of the sample-and-hold circuit 79, AD converter 36 anderror-signal estimating arithmetic unit 33 is identical with that of thefifth embodiment of FIG. 13 and the same effects are obtained.

(K) 11th Embodiment

FIG. 19 is a diagram showing the structure of a radio apparatusaccording to an 11^(th) embodiment having a function for correcting theamplitude and phase of a reference signal. Components identical withthose of the sixth embodiment of FIG. 13 are designated by likereference characters. This embodiment differs in that correctioncircuits 81 ₁˜81 ₄ are provided instead of the frequency shifters 57₁˜57 ₄ to correct the amplitude and phase of the reference signal insuch a manner that the frequency-multiplexed signal (reference signal),which is output from the combiner 58, will agree with the main signalS_(M) output from the combiner 54. The correction circuits 81 ₁˜81 ₄compare, on a per-carrier basis, signals obtained through frequencyconversion by the frequency converters 53 ₁˜53 ₄ with signals obtainedthrough frequency shifting of the transmit baseband signals x₁(t)˜x₄(t)by digital processing, and exercise control in such a manner that thedifferences between these signals become zero, thereby making thereference signal coincide with the main signal.

FIG. 20 is a diagram showing the structure of the correction circuit 81₁; the other correction circuits 81 ₂˜81 ₄ are similarly constructed. AnAD converter 81 a converts the frequency-converted signal output fromthe frequency converter 53 ₁ to a digital signal, and a multiplier 81 binitially multiplies the transmit signal x₁(t) by exp(jωt) to output asignal obtained by digitally shifting the frequency of the transmitsignal x₁(t). A comparator 81 c detects an amplitude difference ν_(d)and a phase difference φ_(d) between these two input signals and obtainsan error Δν=ν_(d)·exp(jφ_(d)). An averaging circuit 81 d averages theoutput of the comparator 81 c to generate an average error Δν_(avr), acomplex-conjugate signal output unit 81 e generates a complex-conjugatevalue Δν_(avr)* of the average error Δν_(avr), a multiplier 81 fmultiplies exp(jωt) by Δν_(avr)*, and a multiplier 81 multiplies thetransmit signal x₁(t) by Δν_(avr)* ·exp(jωt). By repeating theabove-described control, the amplitude difference ν_(d) and phasedifference φ_(d) between the two signals input to the comparator 81 cwill be reduced to zero.

FIG. 21 is a diagram showing the structure of a correction circuitapplicable to the seventh embodiment of FIG. 15. Components identicalwith those of the correction circuit of FIG. 20 are designated by likereference characters. This circuit differs in that a quadraturedemodulator 81 g and AD converters 81 h, 81 i are provided instead ofthe AD converter 81 a. The quadrature demodulator, 81 g appliesquadrature demodulation processing to the output signal of the frequencyconverter 53 ₁, and the AD converters 81 h, 81 i convert the in-phaseand quadrature components output from the quadrature demodulator todigital signals and input the digital signals to the comparator 81 c.Meanwhile, the multiplier 81 b outputs a signal obtained by digitallyshifting the frequency of the baseband transmit signal x₁(t). Thecomparator 81 c detects the amplitude difference ν_(d) and the phasedifference φ_(d) between these two input signals, and the correctioncircuit subsequently exercises control in such a manner that thedifference between these signals becomes zero, in a manner similar tothat of FIG. 20.

FIG. 22 is a diagram showing the structure of a correction circuitapplicable to the eighth embodiment of FIG. 16. Components identicalwith those of the correction circuit of FIG. 20 are designated by likereference characters. This circuit differs in that a digital quadraturedemodulator 81 j is provided on the output side of the AD converter 81a. The AD converter 81 a converts the output signal of the frequencyconverter 53 ₁ to a digital signal, and the quadrature demodulator 81 japplies quadrature demodulation processing digitally to the outputsignal of the signal of the AD converter and inputs the in-phase andquadrature components of the demodulated signal to the comparator 81.Meanwhile, the multiplier 81 b outputs a signal obtained by digitallyshifting the frequency of the baseband transmit signal x₁(t). Thecomparator 81 c detects the amplitude difference ν_(d) and the phasedifference φ_(d) between these two input signals, and the correctioncircuit subsequently exercises control in such a manner that thedifference between these signals becomes zero, in a manner similar tothat of FIG. 20.

(L) 12^(th) Embodiment

FIG. 23 is a diagram showing the structure of a radio apparatusaccording to an 12^(th) embodiment of the present invention, in whichcomponents identical with those of the sixth embodiment of FIG. 13 aredesignated by like reference characters. This embodiment differs in thatPLL circuits 84 ₀˜84 ₄, which are synchronized to exp(jω₀t),exp(jω₁t)˜exp(jω₄t) generated when frequency is shifted digitally, areprovided and supply local signals of frequencies f₀, f₁˜f₄ used when afrequency conversion is performed in analog fashion. In accordance withthe 12^(th) embodiment, the frequencies of an analogfrequency-multiplexed signal and digital frequency-matched signal can besynchronized, thereby making it possible to raise the precision ofdistortion compensation.

(M) 13^(th) Embodiment

FIG. 24 is a diagram showing the structure of a radio apparatusaccording to an 13^(th) embodiment of the present invention, in whichcomponents identical with those of the sixth embodiment of FIG. 13 aredesignated by like reference characters. This embodiment differs in thatDA converters 85 ₁˜85 ₄ are provided and generate local signals offrequencies f₀, f₁˜f₄ by DA-converting exp(jω₁t)˜exp(jω₄t) generatedwhen frequency is shifted digitally. In accordance with the 13^(th)embodiment, the frequencies of an analog frequency-multiplexed signaland digital frequency-matched signal can be synchronized, thereby makingit possible to raise the precision of distortion compensation. It shouldbe noted that the local signal of frequency f₀ is generated by the PLLcircuit 84 ₀ synchronized to exp(jω₀t) in accordance with the 12^(th)embodiment.

(N) 14^(th) Embodiment

FIG. 25 is a diagram showing the structure of a radio apparatusaccording to a 14^(th) embodiment of the present invention. Thisillustrates an example of a case where four frequencies are multiplexed.Components identical with those of the sixth embodiment of FIG. 13 aredesignated by like reference characters. This embodiment differs fromthe sixth embodiment in that frequency shift processing is applieddigitally to the transmit baseband signals x₁(t), x₂(t), X₃(t), x₄(t)and the signals obtained are DA-converted and combined to therebygenerate an analog frequency-multiplexed signal.

The transmit baseband signals x₁(t), x₂(t), X₃(t), x₄(t) are subjectedto a frequency shift by being multiplied by exp(jω₁t), exp(jω₂t),exp(jω₃t), exp(jω₄t) (ω_(n)=2πf_(n)) decided by a desired carrierfrequency spacing, after which the shifted signals are converted toanalog signals by DA converters 51 ₁˜51 ₄ and the combiner 54 combinesthe outputs of these DA converters to generate an analogfrequency-multiplexed signal.

The combiner 32 combines the frequency-multiplexed signal (main signal)S_(M) with the error signal S_(E) output from the error-signalestimating arithmetic unit 33 and inputs the combined signal to thepower amplifier (amp) 34. The distributor 35 branches part of theamplifier output and the frequency converter 55 multiplies the amplifieroutput signal by the local signal of frequency f₀ to down-convert thefrequency and input the resulting signal to the AD converter 36 via thefilter 56. The AD converter 36 AD-converts the input signal and inputsthe obtained digital signal to the error-signal estimating arithmeticunit 33 as the feedback signal S_(F).

Meanwhile, the combiner 58 generates the digital frequency-multiplexedsignal by combining the transmit signals x₁(t)·exp(jω₁t),x₂(t)·exp(jω₂t), x₃(t)·exp(jω₃t), x₄(t)·exp(jω₄t) output from thedigital frequency shifters 57 ₁˜57 ₄. This digital frequency-multiplexedsignal is a signal equivalent to the above-mentioned analogfrequency-multiplexed signal. The frequency shifter 59 subsequentlymultiplies the digital frequency-multiplexed signal by exp(−jω₀t) todown-convert the frequency and input the resulting signal to theerror-signal estimating arithmetic unit 33 as the reference signalS_(R).

The error-signal estimating arithmetic unit 33 uses the reference signalS_(R) and the feedback signal S_(F) to calculate an error signalascribable to non-linear distortion of the amplifier. The DA converter37 DA-converts the obtained error signal and inputs the analog signal tothe frequency converter 61 via the filter 60. The frequency converter 61multiplies the error signal by the signal of frequency f₀ to therebyup-convert the error signal frequency. The attenuator 38 adjusts thelevel of the error signal and then the combiner 32 combines thelevel-adjusted error signal and the main signal. As a result, there isobtained a frequency-multiplexed signal that is the result of adding ona characteristic that is the inverse of the non-linear distortion of theamplifier.

(P) 15^(th) Embodiment

FIG. 26 is a diagram showing the structure of a radio apparatusaccording to a 15^(th) embodiment of the present invention. Thisillustrates an example of a case where four frequencies are multiplexed.Components identical with those of the sixth embodiment of FIG. 13 aredesignated by like reference characters. This embodiment differs fromthe sixth embodiment in that an analog frequency multiplexer AFMP isconstituted by sampling converters 91 ₁˜91 ₄, digital modulator 92 ₁˜92₄, DA converters 51 ₁˜51 ₄ and combiner 54 for combining the outputs ofthese DA converters.

The sampling converters 91 ₁˜91 ₄ convert the rates of the transmitbaseband signals x₁(t), x₂(t), x₃(t), x₄(t) to a sampling rate decidedby carrier frequency. More specifically, the conversion is made to arate that is four times that of the IF carrier frequency output. Afterthe rate conversion is carried out, the digital quadrature modulators 92₁˜92 ₄ digitally modulate the rate-converted signals on a per-carrierbasis and input the modulated signals to the DA converters 51 ₁˜51 ₄.The latter DA-convert the modulated signals to analog signals and thecombiner 54 combines the outputs of the DA converters to generate theanalog frequency-multiplexed signal (main signal) S_(M).

The combiner 32 combines the frequency-multiplexed signal (main signal)with an error signal generated by means, described below, and inputs thecombined signal to the amplifier 34. The distributor 35 branches part ofthe output of amplifier 34 and the frequency converter 55 multiplies theamplifier output signal by the local signal of frequency f₀ todown-convert the frequency and input the resulting signal to the ADconverter 36 via the filter 56. The AD converter 36 AD-converts theinput signal and inputs the obtained digital signal to the error-signalestimating arithmetic unit 33 as the feedback signal S_(F).

Meanwhile, the combiner 58 generates the digital frequency-multiplexedsignal by combining the transmit signals x₁(t)·exp(jω₁t),x₂(t)·exp(jω₂t), x₃(t)·exp(jω₃t), x₄(t)·exp(jω₄t) output from thedigital frequency shifters 57 ₁˜57 ₄. This digital frequency-multiplexedsignal is a signal equivalent to the above-mentioned analogfrequency-multiplexed signal S_(M). The frequency shifter 59subsequently multiplies the digital frequency-multiplexed signal byexp(−jω₀t) to down-convert the frequency and input the resulting signalto the error-signal estimating arithmetic unit 33 as the referencesignal S_(R).

The error-signal estimating arithmetic unit 33 uses the reference signalS_(R) and the feedback signal S_(F) to calculate an error signalascribable to non-linear distortion of the amplifier. The DA converter37 DA-converts the obtained error signal and inputs the analog signal tothe frequency converter 61 via the filter 60. The frequency converter 61multiplies the error signal by the signal of frequency f₀ to therebyup-convert the error signal frequency. The attenuator 38 adjusts thelevel of the error signal and then the combiner 32 combines thelevel-adjusted error signal and the main signal. As a result, there isobtained a frequency-multiplexed signal that is the result of adding ona characteristic that is the inverse of the non-linear distortion of theamplifier.

FIG. 27 is a diagram showing the principles of a sampling-rateconverter. A sampling-rate converter for converting the information rateof a digital signal to a rate multiplied by m/n is constituted by aninterpolator 91 a for inserting (m−1)-number of “0” between input samplesignals, a filter 91 b for dealiasing and a decimator 91 c forextracting values of the necessary sample timing from the filter-outputsample sequence every other n. By virtue of such an m/n sampling rateconverter, a baseband signal x(t) generated at a sampling rate that isthe chip rate or a whole-number multiple of the chip rate is convertedto a signal having a sampling rate decided by the carrier spacing.

FIG. 28 shows an embodiment of the sampling-rate converter. Thesampling-rate converter is implemented by a FIR filter that changes overthe weighting coefficients of filter taps. The FIR filter has aplurality of delay units DLY, a plurality of coefficient multipliersMPL, an adder ADD and a tap controller TCC for changing over the tapcoefficients. The operation performed by the FIR filter is indicated bythe following equation: $\begin{matrix}{{y(i)} = {\sum\limits_{k = {{- {Ntap}}/2}}^{{{Ntap}/2} - 1}\quad {{h\left\lbrack {{mk} + ({ni})_{m}} \right\rbrack}{x\left( {\left\lbrack \frac{ni}{m} \right\rbrack - k} \right)}}}} & (4)\end{matrix}$

(n)_(m)=n modulo m

[a]: largest integer that does not exceed a

Here x(i) represents a discrete time signal at the filter input, y(i) adiscrete time signal at the filter output, and h(k) an impulse responseof the dealiasing filter designed based upon over sampling by a factorof m and tap length N_(tap). For example, with regard to a case wherem=4, n=3, N_(tap)=8 hold, control for changing over the tap coefficientsh(k) in the arrangement of FIG. 28 is as indicated below. Tapcoefficients h(k) are obtained by changing over 32 (=m×N taps) of h(−16)to h(15) in accordance with the result of Equation (2).

time i=0: h(−16) h(−12) h(−8) h(−4) h(0) h(4) h(8) h(12)

time i=1: h(−13) h(−9) h(−5) h(−1) h(3) h(7) h(11) h(15)

time i=2: h(−14) h(−10) h(−6) h(−2) h(2) h(6) h(10)h(14)

time i=3: h(−15) h(−11) h(−7) h(−3) h(1) h(5) h(9) h(13)

time i=4: h(−16) h(−12) h(−8) h(−4) h(0) h(4) h(8) h(12)

time i=5: h(−13) h(−9) h(−5) h(−1) h(3) h(7) h(11) h(15)

Conversion to a sampling rate that is multiplied by 4/3 can beimplemented by a FIR filter using these tap coefficients and weightingcoefficient.

Thus, in accordance with the present invention, it is possible to lowerthe bit precision of DA converters and the conversion speed of ADconverters used in compensating for non-linear distortion of a signalhaving a wide band and a large dynamic range.

Further, in accordance with the present invention, a predistorter for aW-CDMA multicarrier signal that has been difficult to realize until nowcan be implemented with the device capabilities of currently existingD/A and A/D converters.

Further, in accordance with the present invention, an auxiliaryamplifier, delay lines and coupler, etc., that were essential forconventional feed-forward distortion compensation schemes are no longerrequired. This makes it possible to improve the overall efficiency of apower amplifier.

What is claimed is:
 1. A radio apparatus for compensating for non-lineardistortion of a transmission power amplifier, characterized by having: afirst DA converter for converting a transmit signal to an analog signaland outputting the analog transmit signal; a transmission poweramplifier for amplifying and transmitting the transmit signal; branchingmeans for branching part of an output signal of said transmission poweramplifier; an AD converter for converting, to a digital signal, theoutput signal of the transmission power amplifier branched by thebranching means or a signal obtained by subjecting this output signal topredetermined processing, and outputting the digital signal as afeedback signal; an error estimating arithmetic unit for estimating andoutputting an error signal, which is ascribable to non-linear distortionof the amplifier, using the feedback signal and a reference signal,which is the transmit signal; a second DA converter for converting saiderror signal to an analog signal and outputting the analog signal; and acombiner for combining, and outputting to the transmission poweramplifier, the transmit signal converted to the analog signal by thefirst DA converter and the error signal converted to the analog signalby the second DA converter.
 2. A radio apparatus for compensating fornon-linear distortion of a transmission power amplifier which amplifiesand transmits a multicarrier signal for carrying multiple transmitsignals, characterized by having: a first DA converter for convertingeach digital transmit signal to an analog transmit baseband signal andoutputting the analog baseband signal; an analog frequency multiplexerfor subjecting each transmit baseband signal to a frequency-shiftoperation decided by carrier spacing and frequency-multiplexing thefrequency-shifted signals; a transmission power amplifier for amplifyingand transmitting the frequency-multiplexed signal; branching means forbranching part of an output signal of said transmission power amplifier;an AD converter for converting, to a digital signal, the output signalof the transmission power amplifier branched by the branching means or asignal obtained by subjecting this output signal to predeterminedprocessing, and outputting the digital signal as a feedback signal; adigital frequency multiplexer for subjecting each digital transmitsignal digitally to a frequency-shift operation decided by carrierspacing, frequency-multiplexing the frequency-shifted signals andoutputting the frequency-multiplexed signal as a reference signal; anerror estimating arithmetic unit for estimating and outputting an errorsignal, which is ascribable to non-linear distortion of the amplifier,using the reference signal and the feedback signal; a second DAconverter for converting said error signal to an analog signal andoutputting the analog signal; and a combiner for combining, andoutputting to the transmission power amplifier, the error signalconverted to the analog signal by said DA converter and an output signalof said analog frequency multiplexer.
 3. A radio apparatus according toclaim 1, characterized in that said error estimating arithmetic unitobtains error signals by a digital Cartesian scheme in such a mannerthat a difference between said reference signal and said feedback signalbecomes zero, stores said error signals in a memory in correspondencewith transmit-signal values, reads an error signal that corresponds to apresent transmit signal out of the memory and outputs the error signal.4. A radio apparatus according to claim 1, characterized in that saiderror estimating arithmetic unit transforms said reference signal andsaid feedback signal from a rectangular coordinate system to a polarcoordinate system, obtains error signals in such a manner that adifference between said reference signal and said feedback signalbecomes zero in the polar coordinate system, stores the error signals inmemory in correspondence with transmit-signal values, reads an errorsignal that corresponds to a present transmit signal out of the memory,transforms said error signal from the polar coordinate system to therectangular coordinate system and outputs the error signal.
 5. A radioapparatus according to claim 1, characterized in that said errorestimating arithmetic unit obtains distortion compensation coefficientsby adaptive signal processing using said reference signal and saidfeedback signal, stores said distortion compensation coefficients in amemory in correspondence with transmit-signal power values, reads adistortion compensation coefficient corresponding to a power value of apresent transmit signal out of the memory, multiplies said transmitvalue by the distortion compensation coefficient and outputs, as anerror signal, a difference between a transmit signal beforemultiplication by said distortion compensation coefficient and thetransmit signal after multiplication by the distortion compensationcoefficient.
 6. A radio apparatus according to claim 1, characterized inthat said error estimating arithmetic unit obtains, and store in amemory, distortion compensation coefficients by adaptive signalprocessing using said reference signal and said feedback signal, reads adistortion compensation coefficient corresponding to a presenttransmission power value and a past transmit-signal value out of thememory, multiplies the present transmit value by the distortioncompensation coefficient and outputs, as an error signal, a differencebetween a transmit signal before multiplication by said distortioncompensation coefficient and the transmit signal after multiplication bythe distortion compensation coefficient.
 7. A radio apparatus accordingto claim 1, characterized by further having: means for converting atransmit signal to a quadrature signal having an in-phase component anda quadrature component; DA converters, which construct said first DAconverter, for converting the signal components of the quadrature signalto respective ones of analog signals; a first quadrature modulator, towhich the analog signal components output from respective ones of the DAconverters are input, for subjecting the transmit signal to quadraturemodulation; a quadrature demodulator for subjecting an output signal ofthe transmission power amplifier to quadrature demodulation processing;AD converters, which construct said AD converter, for converting anin-phase component and a quadrature component of aquadrature-demodulated signal output from the quadrature modulator torespective ones of digital signals; DA converters, which construct saidsecond DA converter, for converting an in-phase component and aquadrature component of an error signal output from said error-signalestimating arithmetic unit to respective ones of analog signals; and asecond quadrature modulator, to which the analog signal componentsoutput from respective ones of the DA converters are input, forsubjecting the error signal to quadrature modulation; said error-signalestimating arithmetic unit outputting an error signal in such a mannerthat a difference between the in-phase components and a differencebetween the quadrature components of the quadrature signal beforedistortion compensation and the quadrature demodulated signal willbecome zero, and said combiner combines output signals of the first andsecond quadrature modulators and inputs the combined signal to thetransmission power amplifier.
 8. A radio apparatus according to claim 7,characterized by further having sample-and-hold circuits provided on aninput side of respective ones of the AD converters; sampling rate of ADconversion and sampling band being set independently.
 9. A radioapparatus according to claim 1, characterized by further having: meansfor converting a transmit signal to a quadrature signal having anin-phase component and a quadrature component; a first digitalquadrature modulator, to which the input signal components are input,for subjecting the transmit signal to quadrature modulation andinputting the resulting signal to said first DA converter; a digitalquadrature demodulator for subjecting an output signal of said ADconverter to quadrature demodulation processing; and a second digitalquadrature modulator, to which an in-phase component and a quadraturecomponent of an error signal output from said error-signal estimatingarithmetic unit are input, for subjecting said error signal toquadrature modulation and inputting the resulting signal to said secondDA converter; said error-signal estimating arithmetic unit outputting anerror signal in such a manner that a difference between the in-phasecomponents and a difference between the quadrature components of thequadrature signal before distortion compensation and the quadraturedemodulated signal will become zero.
 10. A radio apparatus according toclaim 9, characterized by further having a sample-and-hold circuitprovided on an input side of the AD converter; sampling rate of ADconversion and sampling band being set independently.
 11. A radioapparatus according to claim 2, characterized by further having: meansfor converting respective ones of transmit signals to quadrature signalseach having an in-phase component and a quadrature component; DAconverters, which construct each of said first DA converters, forconverting an in-phase component and a quadrature component of arespective quadrature signal to respective ones of analog signals; afirst quadrature modulator, for each quadrature signal, to which theanalog in-phase and quadrature components output from corresponding DAconverters are input, for subjecting the transmit signal to quadraturemodulation; a quadrature demodulator for subjecting an output signal ofthe transmission power amplifier, or a signal obtained by subjectingsaid output signal to predetermined processing, to quadraturedemodulation processing; AD converters, which construct said ADconverter, for converting an in-phase component and a quadraturecomponent of the quadrature demodulated signal output from thequadrature demodulator to respective ones of digital signals; DAconverters, which construct said second DA converter, for converting anin-phase component and a quadrature component of an error signal outputfrom said error-signal estimating arithmetic unit to respective ones ofanalog signals; and a second quadrature modulator, to which the signalcomponents of the error signal output from said DA converters are input,for subjecting the error signal to quadrature modulation; saiderror-signal estimating arithmetic unit outputting an error signal insuch a manner that a difference between the in-phase components and adifference between the quadrature components of the quadrature signalbefore distortion compensation and the quadrature demodulated signalwill become zero, and said combiner combines output signals of the firstand second quadrature modulators and inputs the combined signal to thetransmission power amplifier.
 12. A radio apparatus according to claim11, characterized by further having sample-and-hold circuits provided onan input side of respective ones of the AD converters; sampling rate ofAD conversion and sampling band being set independently.
 13. A radioapparatus according to claim 2, characterized by further having: meansfor converting respective ones of transmit signals to quadrature signalseach having an in-phase component and a quadrature component; a firstdigital quadrature modulator, for every quadrature signal, to whichdigital in-phase and quadrature components are input, for subjecting thetransmit signal to quadrature modulation and inputting the resultingsignal to said first DA converter; a digital quadrature demodulator forsubjecting an output signal of said AD converter to quadraturedemodulation processing; and a second digital quadrature modulator, towhich an in-phase component and a quadrature component of an errorsignal output from said error-signal estimating arithmetic unit areinput, for subjecting said error signal to quadrature modulation andinputting the resulting signal to said second DA converter; saiderror-signal estimating arithmetic unit outputting an error signal insuch a manner that a difference between the in-phase components and adifference between the quadrature components of the quadrature signalbefore distortion compensation and the quadrature demodulated signalwill become zero.
 14. A radio apparatus according to claim 9,characterized by further having a sample-and-hold circuit provided on aninput side of said AD converter; sampling rate of AD conversion andsampling band being set independently.
 15. A radio apparatus accordingto 2, characterized by further having a correction circuit for everytransmit signal, said correction circuit having an AD converter forconverting, to a digital signal, an analog signal obtained by afrequency shift performed by said analog frequency multiplexer, andmeans for comparing an output of this AD converter with a signalobtained by digitally subjecting said transmit signal to thefrequency-shift operation, and correcting amplitude and phase of theoutput signal based upon result of the comparison; said digitalfrequency multiplexer combining output signals of respective ones of thecorrection circuits and outputting the combined signal as the referencesignal.
 16. A radio apparatus according to claim 11, characterized byfurther having a correction circuit for every transmit signal, saidcorrection circuit having a quadrature demodulator for subjecting ananalog signal, which is obtained by a frequency shift performed by saidanalog frequency multiplexer, to quadrature demodulation processing, ADconverters for converting, to respective ones of digital signals, anin-phase component and a quadrature component of aquadrature-demodulated signal from the quadrature demodulator, and meansfor comparing outputs of these AD converters with signals obtained bydigitally subjecting said transmit signal to the frequency-shiftoperation, and correcting amplitude and phase of the output signal basedupon result of the comparison; said digital frequency multiplexercombining output signals of respective ones of the correction circuitsand outputting the combined signal as the reference signal.
 17. A radioapparatus according to 13, characterized by further having a correctioncircuit for every transmit signal, said correction circuit having an ADconverter for converting, to a digital signal, an analog signal obtainedby a frequency shift performed by said analog frequency multiplexer, adigital quadrature demodulator for subjecting an output signal of thisAD converter to quadrature demodulation processing, and means forcomparing the quadrature-demodulated signal with a signal obtained bydigitally subjecting said transmit signal to the frequency-shiftoperation, and correcting amplitude and phase of the output signal basedupon result of the comparison; said digital frequency multiplexercombining output signals of respective ones of the correction circuitsand outputting the combined signal as the reference signal.
 18. A radioapparatus according to claim 2, characterized by further having PLLcircuits, to which are input digital frequency signals used to subjectthe transmit signals to the frequency-shift operation in the digitalfrequency multiplexer, for generating carrier frequency signals used inthe frequency shift of said analog frequency multiplexer.
 19. A radioapparatus according to claim 2, characterized by further having DAconverters for DA-converting digital frequency signals used to subjectthe transmit signals to the frequency-shift operation in the digitalfrequency multiplexer, and adopting the analog frequency signals, whichare obtained by the DA conversion, as carrier frequency signals used inthe frequency shift of said analog frequency multiplexer.
 20. A radioapparatus for compensating for non-linear distortion of a transmissionpower amplifier which amplifies and transmits a multicarrier signal forcarrying multiple transmit signals, characterized by having: a frequencyshifter for subjecting each digital transmit signal digitally to afrequency-shift operation decided by carrier spacing; a DA converter forDA-converting each frequency-shifted transmit signal; an analogfrequency multiplexer for frequency-multiplexing outputs of each of theDA converters; a transmission power amplifier for amplifying andtransmitting a transmit signal; branching means for branching part of anoutput signal of said transmission power amplifier; an AD converter forconverting, to a digital signal, the output signal of the transmissionpower amplifier branched by said branching means or a signal obtained bysubjecting this output signal to predetermined processing, andoutputting the digital signal as a feedback signal; a digital frequencymultiplexer for digitally combining the frequency-shifted transmitsignals and outputting a reference signal; an error estimatingarithmetic unit for estimating and outputting an error signal, which isascribable to non-linear distortion of the amplifier, using thereference signal and said feedback signal; a DA converter for convertingsaid error signal to an analog signal and outputting the analog signal;and a combiner for combining, and outputting to the transmission poweramplifier, the error signal converted to the analog signal by said DAconverter and an output signal of said analog frequency multiplexer. 21.A radio apparatus for compensating for non-linear distortion of atransmission power amplifier which amplifies and transmits amulticarrier signal for carrying multiple transmit signals,characterized by having: a sampling-rate converter for converting eachdigital transmit signal to a digital signal having a sampling ratedecided by carrier spacing; a digital quadrature modulator forsubjecting an output signal of each sampling-rate converter toquadrature modulation; a DA converter for converting an output signal ofeach quadrature modulator to an analog signal; first combining means forcombining outputs of the DA converters; a transmission power amplifierfor amplifying and transmitting a transmit signal; branching means forbranching part of an output signal of said transmission power amplifier;an AD converter for converting, to a digital signal, the output signalof the transmission power amplifier branched by said branching means ora signal obtained by subjecting this output signal to predeterminedprocessing, and outputting the digital signal as a feedback signal; adigital frequency multiplexer for performing frequency multiplexing bydigitally subjecting each digital transmit signal to a frequency-shiftoperation decided by carrier spacing, and outputting a reference signal;an error estimating arithmetic unit for estimating and outputting anerror signal, which is ascribable to non-linear distortion of theamplifier, using the reference signal and said feedback signal; a DAconverter for converting said error signal to an analog signal andoutputting the analog signal; and a second combiner for combining, andoutputting to the transmission power amplifier, the error signalconverted to the analog signal by said DA converter and an output signalof said first combiner.